Norton noiseless feedback amp calculation

[Jake]

Jake wrote...

Understood. But not the best idea, IMHO.

Like most things in electronic design there are trade offs. I wanted to be
able to more fully understand the realtionship between Zo, the feedback
ratio and operating current.

My thought was that it had to be be better than a 560 ohm resister in the
drain circuit and amplifier stage with no feed back at all. The objective
was to improve the signal handling capability and dynamic range without
messing up the match by to much.

For the parameters with IC and N I need an equation that provides al least
a ball park figure for Zo. Try as I may I can't see how to solve this.

It's certainly laudable to be careful with every dB of gain while
the signal is still weak, or the signal impedance is high, etc.,
but once you've gone through a healthy gain stage, e.g. 20dB,
you no longer need to save every dB (or even every 6dB) and other
considerations take precedence, such as low distortion, accurate
filter properties, etc. By adding an emitter follower (EF), you
lower the impedance, so you can then precisely control it with a
series resistor. The loss of signal is of no consequence. Make
the stage gain 26dB if it worries you that much.

Gain has its own problems of dynamic range with high Zo with a Vs of 10Volt
as well as IP3 for the stage. Feedback has the advantage of added
linearity reducing other undesired products of ajacent and large signals.

Yes. But not a very good approach, unless you're a high volume
manufacturer desperate to save every part. Here, more is better.


Lucky break.

Exactly, thereby precisely setting Zo = 560 ohms to insure the
filter would have its designer-intended frequency response.

Sure but I'll bet IP3 and overall linearity are considerably worse.

Use the EF with say 2mA for Zo = 12 ohms (and with over 500mV
filter-drive capability), and follow it with a 560-ohm resistor.
That's 572 ohms, close enough! Or follow it with 549 ohms 1%
to be right on the money.

1% resisters of such values are like hens teeth. ;-)

I would be more tempted to run at 10mA ;-) and 2.5 ohms with RE of 470E.
Then Rseries match of 560E. Should Zin/hFE not be added to this?

Sorry I have never been much of a fan of resistive matching for RF unless
absolutely no other better way exists. One needs to achieve a good design
balance for the full spectrum of requirements most times.

Retro fitting to existing circuits has it own problem of never enough space
or remote mounting adding even more problems. But if the transformer
feedback proved inferior I'll give it a try.

Improve, yes, but get you where you want to be, no.

A perfect match is not the only requirement but one of a set involving
noise, dynamic range, large signal handling, linearity....

It takes a considerable amount of time and often one has to think real hard
to find a way of evaluating what you need to know. More frustrating is
perhaps better because I know how but lack the equipment. For messing about
there is no chance of recovering a large outlay on seldom used equipment.

Loading the amplifier's high Zo directly with the filter is a
dangerous game, creating distortion as you unwisely rob it of
the excess loop gain it should be using to keep distortion down.

But a sage with no feedback has no such protection or is the IC current
drive to the resister match drain load sufficient to overcome the reflected
impedences?

I am happy to reduce the gain down to 6 dB or whatever the noise figure is
at the filter input plus 1 if needed because it is far easier to add a good
low noise 50 ohm in/out stage at the antenna input ahead of the preselector
or the preselector input which is 50E.

Emitter follower. Cathode follower (CF) without a filament.

Ok but out of interest and because it has now become a mission I would
still like to know how to calculate the approx Zo for the emitter feedback
circuit given.

I am busy putting the final touches to ALC on my cheap and cheerful Chinese
signal generator and that will allow me to more easily check the filter
under both conditions. I gave up trying to control the tubes large output
with junction fets or cathode degeneration and instead regulated the plate
voltage. Pair of signal diodes + TL431 + 4N35 optocoupler and IRF830 as
the series pass element feeding a 150E resister ahead of the smoothing
capacitor. Rectified and regulated heater voltage was used to supply the
TL431 and opto diode. The 4N35 CE shunts the gate and 15 volt protection
zener driven by a 100K resister by the plate voltage connected to the
drain. Crude but works well once the TL431 is stabilised. Then to modify
my old AVO signal generator with 6J5 tube and figure out how to mount the
stuff. ;-) A BF459 and J310 cascode works fine as a 6J5 replacement.

Only one problem on one range the plate supply current needs to be less
than 1mA@20V and my PFET leakage Idss is that. A BU406 chosen for good
gain does no better. VCE sat of the opto is 0.15V. A 200V darlington may
do it but I don't have one handy.

Regards and thanks for your adviced and insight to the problem.

Peter