Norton noiseless feedback amp calculation

[Jake]

+10v
|
T1 |
|
o 1 o N |
UUU UUUUUUUUUU
| | | |
---| | |----------------- Out
| \R1 | | |
| / \ e / c |
--- \ \ / |
C1--- | ----- |
| | | b |
| |-R2--|-----R3------
| | |
====== In
Gnd

The configuration is much the same as the Norton amplifier with the
exception of the input being the base of the transistor.

Assume Ic = 10 ma or in the case of a dual gate FET Id = 8 ma

Since the {1} feedback winding is connected to the emitter the transformer
{T1} has defined impedance's and gain. This is compounded by the summation
of Ie, Ib and Ic input and feedback summation with Re.

How does one calculate this operating point reflected collector impedance
for the output and the stage gain?

Obviously the output may be taken from a tap on the collector winding ie
1:M:N as well.

Can R1, not decoupled, be used to define the output impedance better? I
would think so but how is Zout calculated and the gain change if any.

A net search only finds the Norton configuration but I think Ulrich Rhode
covered this configuration in an old Ham Radio publication under noiseless
feedback.

Any help will be much appreciated as this is beyond my capability for both
transistor and dual gate FET Gm = 18mS

I wanted to use this to replace the first RF amplifier for a Yaesu FRG7
receiver. A dual gate FET is needed because of direct connection to a
preselector tuned circuit. There is a need to match to the 560 ohm filter
input impedance. It has many other applications as well. (Wideband flat
gain and low noise.) Transistor will be a BFR91A and fet BF966 or similar

Thanks
Peter

 

[Jake]

+10v
|
T1 |
|
o 1 o N |
UUU UUUUUUUUUU
| | | |
---| | |----------------- Out
| \R1 | | |
| / \ e / c |
--- \ \ / |
C1--- | ----- |
| | | b |
| |-R2--|-----R3------
| | |
====== In
Gnd

The configuration is much the same as the Norton amplifier with the
exception of the input being the base of the transistor.

Assume Ic = 10 ma or in the case of a dual gate FET Id = 8 ma

Since the {1} feedback winding is connected to the emitter the transformer
{T1} has defined impedance's and gain. This is compounded by the summation
of Ie, Ib and Ic input and feedback summation with Re.

How does one calculate this operating point reflected collector impedance
for the output and the stage gain?

I was not looking for a discussion to the n'th degree and defined down to
pin point accuracy. That thumbsuck is fine and a whole lot better than
nothing.

Some pointers please, a few words to put me on the right track and save
hours of messing about by being able to ball park the behaviour.

Thanks
Peter

 

[Fred Bloggs]

I was not looking for a discussion to the n'th degree and defined down to
pin point accuracy. That thumbsuck is fine and a whole lot better than
nothing.

Some pointers please, a few words to put me on the right track and save
hours of messing about by being able to ball park the behaviour.

Thanks
Peter

From what I recall of the original Norton article, the main thing is
noiseless input impedance termination in addition to gain setting. The
circuit calculation is straightforward. Break it down into the hybrid-pi
corresponding to your operating point- that would be the operating point
computed from the equivalent DC circuit. What is your question anyway?

 

[Winfield Hill]

Jake (Peter) wrote...
Write out the equations and solve them. Routine feedback.

I was not looking for a discussion to the n'th degree and defined
down to pin point accuracy. That thumbsuck is fine and a whole
lot better than nothing.

Some pointers please, a few words to put me on the right track and
save hours of messing about by being able to ball park the behaviour.

OK. Stop complaining.

Look up Dallas Lankford's articles (Microwave Journal, May 1975)
and patents (3,426,298; 3,624,536; and 3,891,934) on common-base
transformer-feedback (CBTF) amplifiers. Good for low-Z inputs.
http://www.kongsfjord.no/ http://www.kongsfjord.no/dl/dl.htm
http://www.kongsfjord.no/dl/Amplifiers/Ultralinear 2N5109 And 2N3053 Amplifiers.pdf

There's also been some good discussion here on s.e.d. And check
Lepaisant in RSI 63-3 (1992). "Low-noise preamplifier with input
and feedback transformers for low source resistance sensors" by
J. Lepaisant, M. Lam Chok Sing, and D. Bloyet

 

[Jake]

Jake (Peter) wrote...

Write out the equations and solve them. Routine feedback.

If I could I would have and not wasted your valuable time and sarcastic
advice.

I have the patents and these deal with a common base amplifier and feedback
via the transformer. Sorry I just don't have the knowledge to write out
new equations where the transformer is terminated in the emitter impedance
and combine the feedback.

While Re maybe 25/Ima this must be combined with the feedback.

How?

OK. Stop complaining.

Look up Dallas Lankford's articles (Microwave Journal, May 1975)
and patents (3,426,298; 3,624,536; and 3,891,934) on common-base
transformer-feedback (CBTF) amplifiers. Good for low-Z inputs.
http://www.kongsfjord.no/ http://www.kongsfjord.no/dl/dl.htm
http://www.kongsfjord.no/dl/Amplifiers/Ultralinear 2N5109 And 2N3053 Amplifiers.pdf

or Zi in = Zout but my use required a high input impedance. The feed point
of those amplifiers ia also the junction of the emitter winding and Rs
which I have decoupled.

The above deals with the case where Rout is determined by Rin or vice
versa. m*m/(m+n+1)Rs = Rl

The decoupled Rs common emitter is determined by other factors such as Re
but that is not all.

I would also be interested to see what then net effect of additional
degenerative feedback is on the comon emmitter circuit.

There's also been some good discussion here on s.e.d. And check
Lepaisant in RSI 63-3 (1992). "Low-noise preamplifier with input
and feedback transformers for low source resistance sensors" by
J. Lepaisant, M. Lam Chok Sing, and D. Bloyet

Sorry this is Africa..... If it is not available on the net I am not going
to get it in a hurry or without traveling some distance and begging to use
a U library.

 

[Jake]

snip

From what I recall of the original Norton article, the main thing is
noiseless input impedance termination in addition to gain setting. The
circuit calculation is straightforward. Break it down into the hybrid-pi
corresponding to your operating point- that would be the operating point
computed from the equivalent DC circuit. What is your question anyway?

Correct but the original is a common base with signal injection at the
emitter winding and external resister junction. Rin determines Rout.

I have that point decoupled and the signal is injected at the base or
source as the case may be.

The use I wanted was for a SW receiver RF amplifier replacement. The
original design dictates how this will be done. The input is from the
preselector tuned circuit so this can't be loaded or Q will be reduced. A
dual gate FET was originally used with Rd = 560 ohms. The output must
match a filter Zin = 560 ohms. Freq 0.5MHz..30MHz.

It is also such a useful building block that I can think of at least a few
other uses even as an output buffer because the transistor version looks
like low output impedances are quite easy to achieve ie 50 ohms with
reasonable gain.

I have two old valve signal generators that desperately need a 50 ohm
output

It's not for production because I can see that Zo will vary with the
operating point and that would need to be stabilised at the very least.

My apology not all are blessed with a higher education and I am totally
self taught. Hybrid Pi is Greek to me ;-0

Regards
Peter

 

[Leon]

If I could I would have and not wasted your valuable time and sarcastic
advice.

I have the patents and these deal with a common base amplifier and feedback
via the transformer. Sorry I just don't have the knowledge to write out
new equations where the transformer is terminated in the emitter impedance
and combine the feedback.

While Re maybe 25/Ima this must be combined with the feedback.

How?


or Zi in = Zout but my use required a high input impedance. The feed point
of those amplifiers ia also the junction of the emitter winding and Rs
which I have decoupled.

The above deals with the case where Rout is determined by Rin or vice
versa. m*m/(m+n+1)Rs = Rl

The decoupled Rs common emitter is determined by other factors such as Re
but that is not all.

I would also be interested to see what then net effect of additional
degenerative feedback is on the comon emmitter circuit.


Sorry this is Africa..... If it is not available on the net I am not going
to get it in a hurry or without traveling some distance and begging to use
a U library.

Win's been very helpful to you and I can't see any sarcasm in his
response - look it up in a dictionary. He was probably niggled by your
apparent inability to search for relevant stuff for yourself.

Leon

 

[Jake]

Win's been very helpful to you and I can't see any sarcasm in his
response - look it up in a dictionary. He was probably niggled by your
apparent inability to search for relevant stuff for yourself.

If I had not done any research and I knew how to model the circuit I would
not have asked. That much is obvious to anyone who reads my original post.
Win made the incorrect assumption I was being lazy and got the correct
response for it. He also incorrectly assumed I had the skills to remodel
the circuit and I have already stated I did not.

Possibly I was niggled by his inability to read and understand, for that I
apologise. His references that I can obtain all point to the common base
version because I have them already.

The only reference I can remember to the common emitter is that of Ulrich
Rhode 1980s Ham Radio article and I don't have a copy and have no source of
obtaining it.

I would appreciate some help, not be told to do what I have done or have
stated I can't do. All the references I can find deal with the common base
version as detailed in my reply. An available reference to the common
emitter version would be nice or some help in remodelling.

I appreciate that the patent papers show a specific application and show
the math but that is to my mind significantly different (Rin = Rout is not
device and operating point determined) to what I wanted. If I knew how to
rearrange it or change it I would have done so and not wasted everyone's
time. Originally clearly stated.

I have monitored this group for some 20 years and have learnt much from
that. I don't ask questions without at least making a large attempt and
effort to answer it for myself.

All I wanted was a bit of help from those far more knowledgeable than
myself.

Regards
Peter

 

[Fred Bloggs]

snip





Correct but the original is a common base with signal injection at the
emitter winding and external resister junction. Rin determines Rout.

I have that point decoupled and the signal is injected at the base or
source as the case may be.

The use I wanted was for a SW receiver RF amplifier replacement. The
original design dictates how this will be done. The input is from the
preselector tuned circuit so this can't be loaded or Q will be reduced. A
dual gate FET was originally used with Rd = 560 ohms. The output must
match a filter Zin = 560 ohms. Freq 0.5MHz..30MHz.

It is also such a useful building block that I can think of at least a few
other uses even as an output buffer because the transistor version looks
like low output impedances are quite easy to achieve ie 50 ohms with
reasonable gain.

I have two old valve signal generators that desperately need a 50 ohm
output

It's not for production because I can see that Zo will vary with the
operating point and that would need to be stabilised at the very least.

My apology not all are blessed with a higher education and I am totally
self taught. Hybrid Pi is Greek to me ;-0

Regards
Peter

http://ece-www.colorado.edu/~bart/book/book/chapter5/ch5_6.htm

 

[Winfield Hill]

Jake wrote...

Sorry this is Africa..... If it is not available on the net I am
not going to get it in a hurry or without traveling some distance
and begging to use a U library.

Well, that what the rest of us have to do. It always amazes
me how some folks who are asking for our help, and want us to
spend our time giving it, can be so demanding, with details
of what they are or are not willing to do on their part.

Anyway, I've added the article, Lepaisant_preamp-xfmrs_RSI.pdf
on my wesbsite, at http://www.picovolt.com/win/elec/articles/

As for the technique, it's simply substituting a transformer,
with its noiseless turns ratio, for resistors in a feedback
circuit. You can use the technique for all kinds of circuits,
but its best value as a low-noise technique comes when you're
working with very low impedances. That's because if resistors
are used, they must have absurdly-low values. For example, if
you're making a 100 pV/root-Hz amplifier, you may need to use
feedback and bias-setting resistors, etc, with values that are
under 0.6 ohms, to keep their Johnson noise from ruining your
amplifier. On the other hand, if you use a transformer, you
just have to keep the winding resistance well under 0.6 ohms.

OTOH, if you're making a high-impedance amplifier, then it's
likely you won't need the transformer's low noise-resistance.

 

[Winfield Hill]

Winfield Hill wrote...

Jake (Peter) wrote...

[ snip ]

There's also been some good discussion here on s.e.d.

Now I remember, check out the s.e.d. discussions about
Jeroen Belleman's amplifier designs, and his web page,
http://jeroen.home.cern.ch/jeroen/tfpu/LNA.shtml

 

[Jake]

From: Winfield Hill - view profile

Date: Sat, Jun 3 2006 5:20 am
Email: Winfield Hill <[email protected]>
Groups: sci.electronics.design

Write out the equations and solve them. Routine feedback.

The feeback is Vo/N
The sum is Vi-Vo/N
-Vo is Hfe(Vi-Vo/N)
Vo/Vi = Hfe/(1+Hfe/N)

So for N = 10:1 and Hfe = 60
Vo/Vi = 8.57

Look up Dallas Lankford's articles (Microwave Journal, May 1975)
and patents (3,426,298; 3,624,536; and 3,891,934) on common-base
transformer-feedback (CBTF) amplifiers. Good for low-Z inputs.
http://www.kongsfjord.no/ http://www.kongsfjord.no/dl/dl.htm
http://www.kongsfjord.no/dl/Amplifiers/Ultralinear 2N5109 And 2...

Not quite applicable as can be seen by the difference in the circuit shown

Zo above is determined almost completely by the impedence seen by the
feedback winding.

There's also been some good discussion here on s.e.d. And check
Lepaisant in RSI 63-3 (1992). "Low-noise preamplifier with input
and feedback transformers for low source resistance sensors" by
J. Lepaisant, M. Lam Chok Sing, and D. Bloyet

Correct this is essentially the same but no treatment is given for
determining Zo.

Without feed back a signal injected at the feedback point would see Re/10mA
Thus Zo is reduced from N*N*Re/Ic(mA) by the feedback
Or for the figures I gave Zo < 100*2.5 = 2K5 ohms

Intuition says its closer to 50..60 ohms.

I can't figure how to calculate it.

Regards and thanks

Peter

 

[Winfield Hill]

Jake wrote...

The feeback is Vo/N
The sum is Vi-Vo/N
-Vo is Hfe(Vi-Vo/N)
Vo/Vi = Hfe/(1+Hfe/N)

So for N = 10:1 and Hfe = 60
Vo/Vi = 8.57


Not quite applicable as can be seen by the difference in the circuit shown

Zo above is determined almost completely by the impedence seen by the
feedback winding.


Correct this is essentially the same but no treatment is given for
determining Zo.

Without feed back a signal injected at the feedback point would
see Re/10mA Thus Zo is reduced from N*N*Re/Ic(mA) by the feedback
Or for the figures I gave Zo < 100*2.5 = 2K5 ohms

Re/10mA, what's that? Do you mean V_T/10mA?

If you were using a BJT, where re = kT/qIc = 2.5 ohms
at 10mA, there are several ways to show Zo = N^2 higher,
or 250 ohms. I measured 150 ohms in a spice analysis,
using a bc547. For a FET, use 1/gm at the operating
current. Zo will likely be 5 to 20x higher, or as high
as 2.5k, etc. You may want to add an emitter follower.

Intuition says its closer to 50..60 ohms.

Explain your intuition.

I can't figure how to calculate it.

You're on the right track if you use 1/gm, rather than hfe
in your thinking. hfe has little effect other than Zin.

A month has gone by, why not make one and measure it?

BTW, did you get all the articles, patents, and web-links
that I referenced, and even posted for you on my website
last June 3rd in this thread? Any thanks due? You think?

 

[Jake]

Jake wrote...

Eek that should be 250 ohms. My apology.

Re/10mA, what's that? Do you mean V_T/10mA?

Sorry I see I made several typo mistakes, should just be Re derived from
the 10mA IC ie 2.5 ohms.

If you were using a BJT, where re = kT/qIc = 2.5 ohms
at 10mA, there are several ways to show Zo = N^2 higher,
or 250 ohms. I measured 150 ohms in a spice analysis,
using a bc547. For a FET, use 1/gm at the operating
current. Zo will likely be 5 to 20x higher, or as high
as 2.5k, etc. You may want to add an emitter follower.

For Zo = 50 ohm with a follower (BJT) implies a IC of 1 mA or less. Is
that a good idea with an amplifier that needs an IP3 of +20..+30dBm?

However in this application I required a Zo = 250 ohms and gain in the
region of 20dB from a FET.

It is possible to tap the output winding at any integer value between 1 and
N. Zo of 450 and 200 could also just as easily be handled by a simple
small additional transmission line transformer on a ferrite bead. In most
cases Ic can be varied by a factor 2 ie 7 to 14mA to acheive a specific Zo.

My original application dictated a minimun of components which had to be
retro fitted to an existing dual gate mosfet RF amplifier feeding a 560 Ohm
Zin filter. Zin had to be high so as not to degrade a resonant tuned
circuit.

But it is an interesting circuit that I wanted to know more about, for
knowledge and who knows.

If Zo can be adjusted by IC then small changes are possible for a one off.
For production it would be a nightmare with unselected FETS.

Low Zin (BJT) can be set with C-B voltage feedback resister. Here Zin
would be Rfb/N.

A test of BFR91A @ 11.4mA and N = 38:3 Emitter resister 56 decoupled by
47nF I measured a Zo of 56 ohms. Vo/Vin = 6. Sorry the frequency was not
noted but it is obvious there was some series feedback via the decoupling
capacitor about 7 ohms worth implying a frequency of about 450KHz.

Explain your intuition.

Gut feel ;-)

You're on the right track if you use 1/gm, rather than hfe
in your thinking. hfe has little effect other than Zin.

OK but how does the feedback influence that? I dont know if I have a
mental block on this but no amount of looking at anything has turned the
light on. All I really want is a better understanding and throwing high
level papers at me will not give that. Consider giving a caveman a match
patent is not going to help him start a fire.

A month has gone by, why not make one and measure it?

I did.

Measure it?

I wanted a ballpark method to cut down on the iterations of winding small
ferrite beads and out of general interest.

I tried a variable resistor (cermit trimmer) across the output winding,
isolated by a capacitor 0,1uF and adjusted until I had half the unloaded
voltage swing (600mV) out. Then measured the resistance. Was this what
you had in mind? Vector impedance measuring equiment does not just ly
around and the last I had access to was 20 plus years ago.

For the FET I got 560 ohms or close enough. Device was a BF966 @ 7mA and N
was 35:3 after several iterations and a check for low frequency response.
The bead used for T1 was an unknown gathered somewhere in a past life.
Initial tests used a Mullard FX1115 bead. Most wound with twisted bifilar
or trifilar 0.198mm wire.

I got caught by the source resister decoupling and not realising the very
low impedences to be expected. That wasted some time and effort in
thinking it was the transformer material and lack of inductance. The
FX1115 would have done fine.

Installing in the receiver (FRG7) also produced an oscillation due to poor
supply line decoupling and layout.

BTW, did you get all the articles, patents, and web-links
that I referenced, and even posted for you on my website
last June 3rd in this thread? Any thanks due? You think?

"Low-noise preamplifier with input and feedback transformers for low source
resistance sensors" by J. Lepaisant, M. Lam Chok Sing, and D. Bloyet was
available on subscription only, thanks for a copy but once again the input
is via the feedback winding in common base configuration so Zo depends on
Zin. This is not the case for the common emitter version with emitter
feedback as shown above, please look again. It may seem obvious to you how
to derive Zo from common base for the common emmitter version but it is not
to me. You can point me in the direction of 1000 papers on the common base
version and it will not help one little bit. If it's to much trouble to
state what one has said was trivial just use a hybrid pi model, then just
say so. I did try and it is beyond a gap I can't seem to cross.

I had all the patents and what was available on the net which deal with the
common base version only. Thanks for the pointer to the SED discussion and
the paper you so kindly made available. Ulrich Rhode gives some detail on
the three configurations in push pull but not any way of determining Zo or
Zin. Yes I found my copy of his Ham Radio article just discussing the
configurations and possible applications after much digging around in 20
year old notes. Wes Haywood also had some stuff published in ARRL but once
again this is based on the patents and common base version.

I wonder how insensitive you would feel knowing my father had passed away.
Thanks for your help, effort and time it is much appreciated.

Peter

 

[Winfield Hill]

Jake wrote...

Winfield wrote,

For Zo = 50 ohm with a follower (BJT) implies a IC of 1 mA or less.
Is that a good idea with an amplifier that needs an IP3 of +20..+30dBm?

However in this application I required a Zo = 250 ohms and gain
in the region of 20dB from a FET.

It's not a good idea to use the output impedance of an emitter
follower as a matching source impedance for a filter, etc.,
because the output impedance is a small-signal value that can
be highly nonlinear with signal swing and transistor current.
Instead establish a nice low Ro value, with 2 to 4mA, etc.,
and then add a series resistor to drive your filter.

For the FET I got 560 ohms or close enough. Device was a BF966 @ 7mA
and N was 35:3 after several iterations and a check for low frequency
response.

Yes, that's reasonable. Add the EF, you'll be happy, then the
exact Zo of the amplifier stage won't matter.

I wonder how insensitive you would feel knowing my father had
passed away. Thanks for your help, effort and time it is much
appreciated.

I'm sorry to hear of your father. It took me several years to
get over mine dying, and I was surprised to discover later that
I had as a result made a direct change just then in some of the
things I had previously enjoyed. I also had unexpected health
responses. It's important to try to maintain a good attitude.

 

[Jake]

Jake wrote...

It's not a good idea to use the output impedance of an emitter
follower as a matching source impedance for a filter, etc.,

Not something I would usually do. I was trying to establish a Zo of 560
ohms in the drain circuit via the feeback winding in the source.

That is another typo above. Sorry.

I was in fact trying to avoid the resistive loss <GB> which would have to
be added to the filter insertion loss. Since it is the RF front end, noise
figure was also important as well as large signal handling capability over
0.5 to 30Mhz.

because the output impedance is a small-signal value that can
be highly nonlinear with signal swing and transistor current.
Instead establish a nice low Ro value, with 2 to 4mA, etc.,
and then add a series resistor to drive your filter.

N turns are the drain winding but Zo must be determined to a large degree
by the source winding which sees via feedback the transformed source
impedance.

I wanted a insight to the calculation of the drain Zo with source feedback
transformer so I could see how well established it was and what would have
the greatest influence on it.

I still have no idea how well established Zo = 560 ohms was for small and
large signals. The original circuit used a 560 ohm drain resister and no
feedback. Unfortunately I don't have equipment to check the filter
response with both amplifier circuits or I would have done this. The
possible influence of the filter reflected impedance changes on the
amplifier drain are also an unknown and my thinking was feedback may
improve this.

Yes, that's reasonable. Add the EF, you'll be happy, then the
exact Zo of the amplifier stage won't matter.

What does EF refer to? Sorry I am not with you on this.

I'm sorry to hear of your father. It took me several years to
get over mine dying, and I was surprised to discover later that
I had as a result made a direct change just then in some of the
things I had previously enjoyed. I also had unexpected health
responses. It's important to try to maintain a good attitude.

Pops was 91 and has a good innings. He died in his sleep at my home where
he has been staying for the last 5 years since my mother died. Gave up
driving at 86 <BG> when he came to stay with me.

61 years is a long time to know somebody and he is very much missed. I
still look around to see why he is not sitting in his favourite places.

Thanks for your kind words and help.

Peter

 

[Jake]

Jake wrote...

It's not a good idea to use the output impedance of an emitter
follower as a matching source impedance for a filter, etc.,

Not something I would usually do. I was trying to establish a Zo of 560
ohms in the drain circuit via the feeback winding in the source.

That is another typo above. Sorry.

I was in fact trying to avoid the resistive loss <GB> which would have to
be added to the filter insertion loss. Since it is the RF front end, noise
figure was also important as well as large signal handling capability over
0.5 to 30Mhz.

because the output impedance is a small-signal value that can
be highly nonlinear with signal swing and transistor current.
Instead establish a nice low Ro value, with 2 to 4mA, etc.,
and then add a series resistor to drive your filter.

N turns are the drain winding but Zo must be determined to a large degree
by the source winding which sees via feedback the transformed source
impedance.

I wanted a insight to the calculation of the drain Zo with source feedback
transformer so I could see how well established it was and what would have
the greatest influence on it.

I still have no idea how well established Zo = 560 ohms was for small and
large signals. The original circuit used a 560 ohm drain resister and no
feedback. Unfortunately I don't have equipment to check the filter
response with both amplifier circuits or I would have done this. The
possible influence of the filter reflected impedance changes on the
amplifier drain are also an unknown and my thinking was feedback may
improve this.

Yes, that's reasonable. Add the EF, you'll be happy, then the
exact Zo of the amplifier stage won't matter.

What does EF refer to? Sorry I am not with you on this.

I'm sorry to hear of your father. It took me several years to
get over mine dying, and I was surprised to discover later that
I had as a result made a direct change just then in some of the
things I had previously enjoyed. I also had unexpected health
responses. It's important to try to maintain a good attitude.

Pops was 91 and has a good innings. He died in his sleep at my home where
he has been staying for the last 5 years since my mother died. Gave up
driving at 86 <BG> when he came to stay with me.

61 years is a long time to know somebody and he is very much missed. I
still look around to see why he is not sitting in his favourite places.

Thanks for your kind words and help.

Peter

 

[Winfield Hill]

Jake wrote...

Not something I would usually do. I was trying to establish a Zo of
560 ohms in the drain circuit via the feeback winding in the source.

Understood. But not the best idea, IMHO.

I was in fact trying to avoid the resistive loss <GB> which would have
to be added to the filter insertion loss. Since it is the RF front end,
noise figure was also important as well as large signal handling
capability over 0.5 to 30Mhz.

It's certainly laudable to be careful with every dB of gain while
the signal is still weak, or the signal impedance is high, etc.,
but once you've gone through a healthy gain stage, e.g. 20dB,
you no longer need to save every dB (or even every 6dB) and other
considerations take precedence, such as low distortion, accurate
filter properties, etc. By adding an emitter follower (EF), you
lower the impedance, so you can then precisely control it with a
series resistor. The loss of signal is of no consequence. Make
the stage gain 26dB if it worries you that much.

N turns are the drain winding but Zo must be determined to a large
degree by the source winding which sees via feedback the transformed
source impedance.
Understood.

I wanted a insight to the calculation of the drain Zo with source
feedback transformer so I could see how well established it was
and what would have the greatest influence on it.

Yes. But not a very good approach, unless you're a high volume
manufacturer desperate to save every part. Here, more is better.

Lucky break.

I still have no idea how well established Zo = 560 ohms was for small
and large signals. The original circuit used a 560 ohm drain resister
and no feedback.

Exactly, thereby precisely setting Zo = 560 ohms to insure the
filter would have its designer-intended frequency response.

Use the EF with say 2mA for Zo = 12 ohms (and with over 500mV
filter-drive capability), and follow it with a 560-ohm resistor.
That's 572 ohms, close enough! Or follow it with 549 ohms 1%
to be right on the money.

Unfortunately I don't have equipment to check the filter response
with both amplifier circuits or I would have done this. The
possible influence of the filter reflected impedance changes on
the amplifier drain are also an unknown and my thinking was feedback
may improve this.

Improve, yes, but get you where you want to be, no.

Loading the amplifier's high Zo directly with the filter is a
dangerous game, creating distortion as you unwisely rob it of
the excess loop gain it should be using to keep distortion down.

What does EF refer to? Sorry I am not with you on this.

Emitter follower. Cathode follower (CF) without a filament.

 

[Jake]

Jake wrote...

Understood. But not the best idea, IMHO.

Like most things in electronic design there are trade offs. I wanted to be
able to more fully understand the realtionship between Zo, the feedback
ratio and operating current.

My thought was that it had to be be better than a 560 ohm resister in the
drain circuit and amplifier stage with no feed back at all. The objective
was to improve the signal handling capability and dynamic range without
messing up the match by to much.

For the parameters with IC and N I need an equation that provides al least
a ball park figure for Zo. Try as I may I can't see how to solve this.

It's certainly laudable to be careful with every dB of gain while
the signal is still weak, or the signal impedance is high, etc.,
but once you've gone through a healthy gain stage, e.g. 20dB,
you no longer need to save every dB (or even every 6dB) and other
considerations take precedence, such as low distortion, accurate
filter properties, etc. By adding an emitter follower (EF), you
lower the impedance, so you can then precisely control it with a
series resistor. The loss of signal is of no consequence. Make
the stage gain 26dB if it worries you that much.

Gain has its own problems of dynamic range with high Zo with a Vs of 10Volt
as well as IP3 for the stage. Feedback has the advantage of added
linearity reducing other undesired products of ajacent and large signals.

Yes. But not a very good approach, unless you're a high volume
manufacturer desperate to save every part. Here, more is better.


Lucky break.

Exactly, thereby precisely setting Zo = 560 ohms to insure the
filter would have its designer-intended frequency response.

Sure but I'll bet IP3 and overall linearity are considerably worse.

Use the EF with say 2mA for Zo = 12 ohms (and with over 500mV
filter-drive capability), and follow it with a 560-ohm resistor.
That's 572 ohms, close enough! Or follow it with 549 ohms 1%
to be right on the money.

1% resisters of such values are like hens teeth. ;-)

I would be more tempted to run at 10mA ;-) and 2.5 ohms with RE of 470E.
Then Rseries match of 560E. Should Zin/hFE not be added to this?

Sorry I have never been much of a fan of resistive matching for RF unless
absolutely no other better way exists. One needs to achieve a good design
balance for the full spectrum of requirements most times.

Retro fitting to existing circuits has it own problem of never enough space
or remote mounting adding even more problems. But if the transformer
feedback proved inferior I'll give it a try.

Improve, yes, but get you where you want to be, no.

A perfect match is not the only requirement but one of a set involving
noise, dynamic range, large signal handling, linearity....

It takes a considerable amount of time and often one has to think real hard
to find a way of evaluating what you need to know. More frustrating is
perhaps better because I know how but lack the equipment. For messing about
there is no chance of recovering a large outlay on seldom used equipment.

Loading the amplifier's high Zo directly with the filter is a
dangerous game, creating distortion as you unwisely rob it of
the excess loop gain it should be using to keep distortion down.

But a sage with no feedback has no such protection or is the IC current
drive to the resister match drain load sufficient to overcome the reflected
impedences?

I am happy to reduce the gain down to 6 dB or whatever the noise figure is
at the filter input plus 1 if needed because it is far easier to add a good
low noise 50 ohm in/out stage at the antenna input ahead of the preselector
or the preselector input which is 50E and 1st RF amp.

Emitter follower. Cathode follower (CF) without a filament.

Ok but out of interest and because it has now become a mission I would
still like to know how to calculate the approx Zo for the emitter feedback
circuit given.

I am busy putting the final touches to ALC on my cheap and cheerful Chinese
signal generator and that will allow me to more easily check the filter
under both conditions. I gave up trying to control the tubes large output
with junction fets or cathode degeneration and instead regulated the plate
voltage. Pair of signal diodes + TL431 + 4N35 optocoupler and IRF830 as
the series pass element feeding a 150E resister ahead of the smoothing
capacitor. Rectified and regulated heater voltage was used to supply the
TL431 and opto diode. The 4N35 CE shunts the gate and 15 volt protection
zener driven by a 100K resister by the plate voltage connected to the
drain. Crude but works well once the TL431 is stabilised. Then to modify
my old AVO signal generator with 6J5 tube and figure out how to mount the
stuff. ;-) A BF495 and J310 cascode works fine as a 6J5 replacement.

Regards and thanks for your adviced and insight to the problem.

Peter

 

[Winfield Hill]

Jake wrote...

Winfield Hill wrote:

Sorry I have never been much of a fan of resistive matching for
RF unless absolutely no other better way exists. One needs to
achieve a good design balance for the full spectrum of
requirements most times.

Your design criteria seems to me to harken back to the days of
expensive components and assembly costs, minimal parts count,
and painful "optimization" (which, in many celebrated cases,
absolutely did *not* work out). Not being disrespectful, but
thankfully I left that design approach behind 38 years ago.

But a stage with no feedback has no such protection ...

You're comparing apples and oranges: A common-source stage with
a highly-linear predictable Zo and an uncontrolled-distortion,
with a precisely-controlled low-distortion gain stage with a
fragile "don't-tamper-with-me" Zout. That's how I see it. I
prefer the second choice, because I can easily fix its problems
and thereby achieve its low-distortion low-noise Nirvana.

IMHO, the low-distortion gain stage is the winner, hands down.

Regards and thanks for your adviced and insight to the problem.

You're very welcome. I'm going to retire from the conversation,
because your interests seem to move quite separately from mine,
and I have too many pressing projects on the burner now. By all
means, enjoy yourself. Let us know how your project progresses.