low-cost 1800-amp heating source

[Tony Williams]

It looks like my series case has enough faults to lose out tot he
parallel case. You could feed the control current through an
inductor so that the ripple voltage doesn't get to the control
circuit.

I'll back away from my sweeping statement that when
in paralleled primaries the control current should
be from a high AC impedance.

The reason for that is that the measured AC current
in the 10000uF does not exceed 0.4Arms and that only
adds 128mW to the dissipation in this control winding
(from 3.2W up to 3.328W at 2Adc control current).

Having the 10000uF then makes a PWM'd control current
much easier because the inductor no longer has to
have a high impedance at 50/60Hz.

I have been thinking about using a "light dimmer" style circuit
to make the control current. Since the ripple from noe hits its
maximum just at the zero crossing, it will change the timing of
when the core saturates. In the current design, the cores
saturate late in the alternation. This would shift it even later
making the power factor worse. It would be nice if the control
circuit instead made the power factor better.

[snip]

For 50W of control power into a low resistance load
I suspect it has to be a PWM'd field coil driver type
of circuit.

Vsupply ---------+----------+-----+
| | |
| ) |
| Rsec1) |
_|_ ) |+
/_\ | ===10000uF
| ) |
| Rsec2) |
| ) |
| | |
| +-----+
| |
| |)
| |)Inductor
| |)
| |
+----------+
|--+
PWM--> ------||
|--+
|
<--Ipeak---------+
|
[Rshunt]
|
0v------------------------+------

Vsupply could even be unregulated.

 

[Glen Walpert]

Interesting to see this discussion migrate to the design of
magnetic amplifiers, or mag-amps. [snip]
Typical 1940's application ran the mag-amp control winding at a
fairly high voltage from a tube amp, then ran the DC output from
the mag-amp to the generator field of a big motor-generator set
which provided power to the load. State of the art for fast,
accurate positioning of large objects at the time.

There was a DC output mag-amp called the Ramey Amplifier.
A quick google on that produced on the first page a
very interesting Swedish paper, about controlling the
field of a JAS-39 generator with a 3-phase Ramey amp.
Quite an in-depth treatment, including the calculation
of core losses and photographs of stacked toroidal
inductors with the control winding wound around the
stack. Early 1990's apparently.

Nice paper, thanks.
www.diva-portal.org/diva/getDocument?urn_nbn_se_kth_diva-4439-2__fulltext.pdf

Mag amps are also often used as switches for applications like SCR
gate drive where isolation is required, and apparently for SMPS output
regulation:
www.mag-inc.com/pdf/sr-4.pdf

I think they might be used more if more people understood them. Hey
Win, how about another addition to your next AoE edition? (Said
without checking your exixting book for mag-amp info only because it
seems to have wandered away from the bookshelf where it belongs. My
failure to remember such is hardly conclusive.

Glen

 

[Tony Williams]

Nice paper, thanks.
www.diva-portal.org/diva/getDocument?urn_nbn_se_kth_diva-4439-2__fulltext.pdf

That's him, thanks. About a 4.5meg download.

That paper also discusses Harry's point about the
best magnetic material to use.

 

[MooseFET]

I'll back away from my sweeping statement that when
in paralleled primaries the control current should
be from a high AC impedance.

The reason for that is that the measured AC current
in the 10000uF does not exceed 0.4Arms and that only
adds 128mW to the dissipation in this control winding
(from 3.2W up to 3.328W at 2Adc control current).

I assume you are willing to waste 128mW. I think you may have jumped
from this to the idea that it is better to waste the power. I disagree
as I will go into below.

Having the 10000uF then makes a PWM'd control current
much easier because the inductor no longer has to
have a high impedance at 50/60Hz.

I disagree with this I will explain after the modified drawing.

Your version for reference of those reading:

Vsupply ---------+----------+-----+
| | |
| ) |
| Rsec1) |
_|_ ) |+
/_\ | ===10000uF
| ) |
| Rsec2) |
| ) |
| | |
| +-----+
| |
| |)
| |)Inductor
| |)
| |
+----------+
|--+
PWM--> ------||
|--+
|
<--Ipeak---------+
|
[Rshunt]
|
0v------------------------+------

My version:

----------
Vsupply ---------+----------+-----! Down !
| | ! shifting !---IL
! \ ! circuit !
! / ! !
! \ ! !
! ! ! !
! +-----! !
! ! ----------
| )
| Rsec1)
_|_ )
/_\ |
| )
| Rsec2)
| )
| |
| !
| |
| |)
--!!-- | |)Inductor
! ! | |)
+-/\/\-+ | |
! ! +----------+
Iset-/\/\+-!+\ ! |--+
! >-+---------||
IL--!-/ |--+
|
GND

The real inductor blocks the bulk of the AC from getting to the
control windings.

I loose more power in the sense resistor but it makes the control
circuit much easier. By removing the capacitor on the windings we end
up controlling the actual current in the control windings and not the
voltage on them. This mostly takes the pole from the inductance of
the windings out of the servo loop. The PWM shows a very high
impedance to the control windings.

I assume you have a feedback from the temperature of the heater to the
Iset. This will have a large phase lag. The mass will integrate and
the distance from the heater to the thermistor may add a transport
delay. The last thing you will want is another couple of poles in the
control circuit of the saturable reactor.

Vsupply could even be unregulated.

Yes. Both your and my designs are basically regulators on their own.

 

[Tony Williams]

I assume you are willing to waste 128mW. I think you may have
jumped from this to the idea that it is better to waste the
power. I disagree as I will go into below.

I disagree with this I will explain after the modified drawing.

----------
Vsupply ---------+----------+-----! Down !
| | ! shifting !---IL
! \ ! circuit !
! / ! !
! \ ! !
! ! ! !
! +-----! !
! ! ----------
| )
| Rsec1)
_|_ )
/_\ |
| )
| Rsec2)
| )
| |
| !
| |
| |)
--!!-- | |)Inductor
! ! | |)
+-/\/\-+ | |
! ! +----------+
Iset-/\/\+-!+\ ! |--+
! >-+---------||
IL--!-/ |--+
|
GND

The real inductor blocks the bulk of the AC from getting to the
control windings.

The reason I backed away from a high impedance drive was
the realisation that the bottom end of that inductor is
always at a low impedance point, either 0V or Vsupply.

This means that, irrespective of the PWM frequency, the
inductor has to have a high enough value such that it
presents a high impedance to the 100Hz ripple voltage.
That's 'high' relative to the source impedance of the
ripple voltage.

I did a crude attempt at measurement of the source
impedance, measuring the change in the pk-pk volts
as a result of changes in resistive loading of the
control winding.

At 0.5A, 1A, and 1.5A DC control current, the output
impedance of the ripple was around 1.5 to 1.6 ohms.

The inductor's impedance has to be much higher than
1.5 ohms. The waveshape of the ripple is not a sinewave
it is saturation difference spikes at 100Hz so the
inductance need not be as high as first thought, but
it will probably need to be in the mH region, (at 2A
dc polarising current).

BTW: The source resistance seen by those two primaries is
the 590 ohm load and I am attracted by the coincidence
that 590 ohm transformed by two 240:18V transformers
in parallel would present 1.65 ohms on the secondaries.
That's probably yet another bizarre TW-ism though.

BTW2: The voltage across each secondary is of the order
of 55V pk-pk, and the resultant difference ripple is
around 2.5V pk-pk maximum. So they are not doing too bad
a job of looking at each other whilst one of them is
going into a region of low relative permeability.

I loose more power in the sense resistor but it makes the control
circuit much easier. By removing the capacitor on the windings
we end up controlling the actual current in the control windings
and not the voltage on them. This mostly takes the pole from the
inductance of the windings out of the servo loop. The PWM shows
a very high impedance to the control windings.

I had thought that the transformers effectively shorted
each other out so the impedance looking back into the
control winding would be mostly resistive. Worth a
quick check then.

The input impedance was measured at 50Hz and 5KHz
with a signal generator and scope. Calcs suggest the
control input looks like 1.4 ohms and 92uH in series.

1.4 ohms is near the sum of the secondary and transformed
primary resistances (1.3R), and I suspect that 92uH is
something like two leakage inductances in series.

I assume you have a feedback from the temperature of the heater
to the Iset. This will have a large phase lag. The mass will
integrate and the distance from the heater to the thermistor may
add a transport delay. The last thing you will want is another
couple of poles in the control circuit of the saturable reactor.

That's getting far too complicated for me atm.

 

[MooseFET]

The reason I backed away from a high impedance drive was
the realisation that the bottom end of that inductor is
always at a low impedance point, either 0V or Vsupply.

I think this is an error. Observing my PWM system: If the current in
the induct attempts to change from the desired set point, the
comparitor turns the MOSFET on or off. This means that for low
frequencies and small signals, the impedance is extremely high.

You didn't show what was hooked to your current sense so It may or may
not also be true of your design.

The big question is whether the situation is "small signal" or not.


[....]

I did a crude attempt at measurement of the source
impedance, measuring the change in the pk-pk volts
as a result of changes in resistive loading of the
control winding.

At 0.5A, 1A, and 1.5A DC control current, the output
impedance of the ripple was around 1.5 to 1.6 ohms.

I am slightly surprised that the impedance of it is so high. I had
figured it would be mostly the resistances of the transformers.

[.....]

BTW: The source resistance seen by those two primaries is
the 590 ohm load and I am attracted by the coincidence
that 590 ohm transformed by two 240:18V transformers
in parallel would present 1.65 ohms on the secondaries.
That's probably yet another bizarre TW-ism though.

That may be it. The 100Hz depends on the AC current in the primary
causing saturation. Imagining putting a zero AC impedance between the
control windings, I see that changing the current in the load.

This means that there is a path for the load's impedance to get
translated into something we see at the control side.

BTW2: The voltage across each secondary is of the order
of 55V pk-pk, and the resultant difference ripple is
around 2.5V pk-pk maximum. So they are not doing too bad
a job of looking at each other whilst one of them is
going into a region of low relative permeability.

That is low enough that this may really be "low signal".

I had thought that the transformers effectively shorted
each other out so the impedance looking back into the
control winding would be mostly resistive. Worth a
quick check then.

The input impedance was measured at 50Hz and 5KHz
with a signal generator and scope. Calcs suggest the
control input looks like 1.4 ohms and 92uH in series.

I think this sounds right. That is quite a small time constant so it
is not a problem for your servo design.

1.4 ohms is near the sum of the secondary and transformed
primary resistances (1.3R), and I suspect that 92uH is
something like two leakage inductances in series.


That's getting far too complicated for me atm.

"Press on and the light will shine"

 

[Tony Williams]

The reason I backed away from a high impedance drive was
the realisation that the bottom end of that inductor is
always at a low impedance point, either 0V or Vsupply.

[/QUOTE]

I think this is an error. Observing my PWM system: If the
current in the induct attempts to change from the desired set
point, the comparitor turns the MOSFET on or off. This means
that for low frequencies and small signals, the impedance is
extremely high.

Sorry Ken this has been me being thick and I now
see what you have been saying.

For ref: I had a quick look at the FFT of the
spiky ripple at the control terminals a few days
ago. The harmonics extended out to 5KHz, although
they are generally 40dB down by 2.5KHz. That
gives some idea of the required bandwidth of an
inner current servo loop.

You didn't show what was hooked to your current sense so It may
or may not also be true of your design.

No I was locked in field coil mode, running the
PWM stage in low gain open loop, controlled by
the outer loop. I had not even thought of having
a current-demand inner loop that was fast enough to
take care of the ripple.

snip rest.

 

[MooseFET]

For ref: I had a quick look at the FFT of the
spiky ripple at the control terminals a few days
ago. The harmonics extended out to 5KHz, although
they are generally 40dB down by 2.5KHz. That
gives some idea of the required bandwidth of an
inner current servo loop.

The simple minded comparitor sort of design will work if the ripple
never causes us to get outside the "small signal" case where the pulse
width is varying continuously.

The biggest problem in this area is that the loop can't stop the
current. We could add a second power MOSFET and diode and make it so
that we can tranfer power back onto the supply rail if we want to slow
the current down. This would require a small dead band so that we
aren't constantly engaging that part of the circuit.

If we don't need the extra circuits, the design of the control part is
too simple to be very interesting. Even with it things aren't very
complex. There are two comparitors with slightly different thresholds
and a little positive feedback.

If I was doing it, I would have the positive feedback of each
comparitor also go to the input of the other and have one extra path
in the feedback I will new describe:

With the "increase the current" comparitor goes to the off condition,
a series RC from its output reaches over to the "decrease the current"
comparitor's inputs. This differentiated signal prevents the
"decrease the current" comparitor from acting for a brief time. This
gives the resistance of the control circuit a little time to reduce
the current before the extra MOSFET is activated for the purpose.


This extra MOSFET for the "decrease the current" requires a level
shifting driver. We can use an N channel device if we have the needed
higher supply voltage. This higher voltage only needs to supply
several mA so it wouldn't be hard to come up with.


Higher ----/\/\-----+----+----+-----------------
! ! ! !
/-/ --- ! !/
^ --- --/\/\----+----! NPN
! ! ! !\e
! ! ! ! !!---Vcc
! ! ! +-------!!
! ! ! ! !!---+--
Inductor
! ! ! !/e !
! ! +----! PNP !
! ! ! !\ !
! ! ! ! !
----+----------------------+------------+--!<---
GND
!
!
!!--- Supertex
Off Drive-----------------!! TN0640
!!-----/\/\---GND
R2

The TN0640 provides way more current than the PNP needs when the
inductors voltage is up. R2 must be low enough that we can be sure to
keep the N channel off when the inductor swings one drop below ground.