Maker Pro
Maker Pro

Core power handling Capability

H

Hammy

I have some Flyback transformers (from CoilCraft) these were
originally intended for use in a 70W 60kHz application using the
NCP1200. If I increase the switching frequency to 100kHz is it
reasonable to expect 90-100W using the same core?

I've been running simulations and the rms and peak current for 100kHz
equalizes at 90W ,with the 70W 60kHz flyback. There is a higher DC
component of course (about 10-15%) due to a reduced switching cycle,
the converter is deeper in CCM.

I had a sheet giving approximation on core power throughput vs
frequency and it shows that for 70W 70kHz increasing the Frequency to
100kHz the same E-Core could handle a little over 100W. I know I could
expect higher DC winding losses; in a pinch I could add another
parallel winding there is already two it would be tight though.

Any magnetic experts out there?
 
H

Hammy

The spread-sheet assumes that you are reconfiguring turns, gap and
wire guage for the different operating frequency.

Simply increasing the frequency may reduce flux peaks, but core loss
will not reduce proportionally if the frequency is increased at the
same time. The net result is increased total loss even without an
increased current through-put.

Copper losses increase with the square of the current density.

What actually happens depends on how the original transformer was
designed. If you don't know the core shape, material, turns, wire
gauge, voltage ratios and frequency, there's little point in
speculating.

It is from this application note from On Semi.

http://www.onsemi.com/pub_link/Collateral/AND8076-D.PDF

This is pretty well all they have for the transformer.

"Transformer

Below are the key parameters you will pass to your
transformer manufacturer to help him select the right
winding size and tailor the internal gap:
Maximum peak primary current, including 160 ns
propagation delay: 1 / 0.33 + 374 × 160 n / 700 m = 3.2 A
Maximum primary RMS current at low line: 1.6 A
Maximum secondary RMS current: 6.9 A
Primary inductance: 700 mH
Turn-ratio, power section: Np:Ns = 1:0.166
Turn-ratio, auxiliary section: Np:Naux = 1:0.15"

I'm not exceeding any of those ratings upto 90W with a safety margin.
If I had to guess and I do it may be either a ETD34 its about 1/2 a cm
longer then an ETD29 core that I have or an EE32.
It'd be faster just to fire up the circuit and measure deltaT as the
frequency-setting component is altered - if the circuit is
pre-existing and you nkow no more about it.

Yes your probably right. I will just hook it up and monitor for
saturation and temp rise.
If the circuit is not pre-existing, you'll get better results working
with fewer unknowns. Transformer loss is not the only consideration as
a flyback circuit's throughput power increases.

Components have improved since they did the AN I'm using a FET with
1/5 the rdson with smaller gate charge then the one they used as well
as sync-rectification.That alone reduces the losses compared to
original design by about 4W.
 
L

legg

I have some Flyback transformers (from CoilCraft) these were
originally intended for use in a 70W 60kHz application using the
NCP1200. If I increase the switching frequency to 100kHz is it
reasonable to expect 90-100W using the same core?

I've been running simulations and the rms and peak current for 100kHz
equalizes at 90W ,with the 70W 60kHz flyback. There is a higher DC
component of course (about 10-15%) due to a reduced switching cycle,
the converter is deeper in CCM.

I had a sheet giving approximation on core power throughput vs
frequency and it shows that for 70W 70kHz increasing the Frequency to
100kHz the same E-Core could handle a little over 100W. I know I could
expect higher DC winding losses; in a pinch I could add another
parallel winding there is already two it would be tight though.
The spread-sheet assumes that you are reconfiguring turns, gap and
wire guage for the different operating frequency.

Simply increasing the frequency may reduce flux peaks, but core loss
will not reduce proportionally if the frequency is increased at the
same time. The net result is increased total loss even without an
increased current through-put.

Copper losses increase with the square of the current density.

What actually happens depends on how the original transformer was
designed. If you don't know the core shape, material, turns, wire
gauge, voltage ratios and frequency, there's little point in
speculating.

It'd be faster just to fire up the circuit and measure deltaT as the
frequency-setting component is altered - if the circuit is
pre-existing and you nkow no more about it.

If the circuit is not pre-existing, you'll get better results working
with fewer unknowns. Transformer loss is not the only consideration as
a flyback circuit's throughput power increases.

RL
 
D

Don Klipstein

I have some Flyback transformers (from CoilCraft) these were
originally intended for use in a 70W 60kHz application using the
NCP1200. If I increase the switching frequency to 100kHz is it
reasonable to expect 90-100W using the same core?

I've been running simulations and the rms and peak current for 100kHz
equalizes at 90W ,with the 70W 60kHz flyback. There is a higher DC
component of course (about 10-15%) due to a reduced switching cycle,
the converter is deeper in CCM.

I had a sheet giving approximation on core power throughput vs
frequency and it shows that for 70W 70kHz increasing the Frequency to
100kHz the same E-Core could handle a little over 100W. I know I could
expect higher DC winding losses; in a pinch I could add another
parallel winding there is already two it would be tight though.

Any magnetic experts out there?

The loss in a ferrite core should be mostly hysteresis loss, which is
roughly proportional to frequency and square of magnetic field intensity.
Power throughput, at least in an oversimplified case, is proportional to
the squares of frequency and magnetic field intensity.

Ideally, ratio of throughput to core loss is proportional to frequency.

Meanwhile, there are other issues:

1. Resistance of the copper will be higher at the higher frequency,
approaching proportional to the square root of frequency once the "skin
depth" gets much smaller than the wire radius.

2. The transformer may have enough inter-layer capacitance to cause
a significant lowpass filter effect. Do you know that it will work at the
higher frequency?

Or are you planning to rewind it?

3. If you rewind it and use fewer turns of thicker wire, keep in mind
that the wire's resistance at the frequency in question may be closer to
inverse proportional to the wire's circumference than to its cross section
area due to the skin effect.

4. The switching transistor's switching loss, as a percentage of power
throughput, is proportional to frequency. The transistor's rise and fall
times will probably be in the tens of nanoseconds, possibly more.

Consider the energy dissipated assuming rise time times half the current
initially conducted by the transistor (possibly zero) times the input
voltage,
plus the fall time times half the current being conducted by the
transistor shortly before switch-off times the voltage that the transistor
experiences during switch-off (always more than the input voltage, usually
by a factor of more than 2, in flyback circuits).

Keep in mind that rise and fall times in transistor datasheets are at
junction temperature of 25 degrees C and with ideal or very good driving
of the transistor. In actual applications, these times are usually
slower (longer periods of time).

Multiply the switching losses by frequency, add conduction losses
(increases with temperature if the tyransistor is a power MOSFET), and
determine if that is going to be too much heat for the transistor to
dissipate. If the heatsinking is currently minimal, then it is *probably*
easy to hack additional or greater heatsinking onto the switching
transistor.

- Don Klipstein ([email protected])
 
L

legg

It is from this application note from On Semi.

http://www.onsemi.com/pub_link/Collateral/AND8076-D.PDF

This is pretty well all they have for the transformer.

"Transformer

Below are the key parameters you will pass to your
transformer manufacturer to help him select the right
winding size and tailor the internal gap:
Maximum peak primary current, including 160 ns
propagation delay: 1 / 0.33 + 374 × 160 n / 700 m = 3.2 A
Maximum primary RMS current at low line: 1.6 A
Maximum secondary RMS current: 6.9 A
Primary inductance: 700 mH
Turn-ratio, power section: Np:Ns = 1:0.166
Turn-ratio, auxiliary section: Np:Naux = 1:0.15"

I'm not exceeding any of those ratings upto 90W with a safety margin.
If I had to guess and I do it may be either a ETD34 its about 1/2 a cm
longer then an ETD29 core that I have or an EE32.
Quite frankly, I've never seen a power transformer described in this
manner, nor is the list actually complete without it's surrounding
article, in which topology, operating frequency, input voltage range,
and output voltage are actually mentioned. It has to be assumed that
the part must meet the requirements of some coordinated safety
standard ~ 60950.

The article itself seems a little quirky - focussing more on control
chip pecadiloes than power train. As the part is being made available
for this application, there is little emphasis on practical
transformer design issues.

One example; the choice of primary inductance is made arbitrarily on
the basis of full load transition from complete to incomplete energy
transfer mode (an irrelevant feature) at an arbitrary input voltage,
somewhere (the author hopes) the psu will never actually have to run.
Then this careful calculation is (equally arbitrarily) approximately
doubled.

Another interesting issue is the fact that, in the end, the author
somehow achieves the design spec output power only at a higher output
voltage than intended. Whether this indicates that the final iteration
was incapable of the design spec current and voltage, or that the
author was just to lazy to complete an accurate set of drawings for
his article - is a mystery to me.

You'd be better off haunting the old Unitrode seminar app notes, if
you're interested in flyback transformer design iteration.

http://focus.ti.com/docs/training/catalog/events/event.jhtml?sku=SEM401014

RL
 
H

Hammy

Quite frankly, I've never seen a power transformer described in this
manner, nor is the list actually complete without it's surrounding
article, in which topology, operating frequency, input voltage range,
and output voltage are actually mentioned. It has to be assumed that
the part must meet the requirements of some coordinated safety
standard ~ 60950.

Yes it is from Coilcraft; here;

http://www.coilcraft.com/y8844.cfm

Its the Z9260-AL

I've noticed that some game consoles have the exact same specs PSU
wise. 16.5V, 4.2A. I never took one apart but I'm guessing they are
the PSU from the application note.
The article itself seems a little quirky - focussing more on control
chip pecadiloes than power train. As the part is being made available
for this application, there is little emphasis on practical
transformer design issues.

Well it is written by C.Basso for Onsemi so I view like I view most
application notes like commercials.

http://pagesperso-orange.fr/cbasso/Spice.htm

One example; the choice of primary inductance is made arbitrarily on
the basis of full load transition from complete to incomplete energy
transfer mode (an irrelevant feature) at an arbitrary input voltage,
somewhere (the author hopes) the psu will never actually have to run.
Then this careful calculation is (equally arbitrarily) approximately
doubled.

This I understand why they would do it. I don't think you would want a
Flyback much above 30W operating in DCM at low line peak current could
start to get high.

The reason they give for adjusting the inductance is to reduce the
peak current; the controller enters a burst/ skip mode at one third
full load if the peak current is high the supply would generate
excessive noise. The burst mode is in the audible frequency range.

They probably tested and determined that 700uH is when the supply is
tolerable noise wise.

They could have just as easily decreased the level at which the
controller enters burst mode.
Another interesting issue is the fact that, in the end, the author
somehow achieves the design spec output power only at a higher output
voltage than intended. Whether this indicates that the final iteration
was incapable of the design spec current and voltage, or that the
author was just to lazy to complete an accurate set of drawings for
his article - is a mystery to me.

You noticed that too:) I was curious about why they did that as well.
They mentioned earlier in the AN about having difficulty in
maintaining sufficient voltage level on the AUX winding during burst
mode this might have something to do with it.

They are trying to highlight the amazing low standby power the chip
can achieve. This can only be obtained with the DSS inactive and the
controller being supplied from the aux winding.

I'm hoping to have the time to get a board done up some time through
the week.

I'm using the NCP1217 100kHz version. I found an appnote using that
controller and the same CoilCraft transformer. The power output is 85W
but again it's higher output voltage at 3.5A.

http://www.onsemi.com/pub_link/Collateral/DN06038-D.PDF
 
Top